Removal of ICI/ISI errors in frequency domain channel estimation for wireless repeaters

ABSTRACT

A method for estimating a feedback channel for a wireless repeater using frequency domain channel estimation estimates an error correction term using a most recent channel estimate and cancels the error correction term from a current block of the receive signal. Then, the feedback channel is estimated using frequency domain channel estimation and using a current block of the pilot signal and the corrected block of the receive signal. A channel estimate error term may also be estimated and subtracted directly from the channel estimate.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional PatentApplication Ser. No. 61/177,198, filed on May 11, 2009, whichapplication is incorporated herein by reference in its entirety.

This application is related to the following concurrently filed andcommonly assigned U.S. patent applications: application Ser. No.12/776,786, entitled “Removal of Multiplicative Errors in FrequencyDomain Channel Estimation For Wireless Repeaters”, filed May 11, 2009;application Ser. No. 12/776,867 and now U.S. Pat. No. 8,611,227,entitled “Frequency Domain Feedback Channel Estimation for anInterference Cancellation Repeater Including Sampling of Non CausalTaps”, filed May 11, 2009; and application Ser. No. 12/776,957, entitled“Channel Estimate Pruning in Presence of Large Signal Dynamics in anInterference Cancellation Repeater”, filed May 11, 2014. Theapplications are incorporated herein by reference in their entireties.

BACKGROUND

1. Field

This disclosure generally relates to repeaters in wireless communicationsystems.

2. Background

Wireless communication systems and techniques have become an importantpart of the way we communicate. However, providing coverage can be asignificant challenge to wireless service providers. One way to extendcoverage is to deploy repeaters.

In general, a repeater is a device that receives a signal, amplifies thesignal, and transmits the amplified signal. FIG. 1 shows a basic diagramof a repeater 110, in the context of a cellular telephone system.Repeater 110 includes a donor antenna 115 as an example networkinterface to network infrastructure such as a base station 125. Repeater110 also includes a server antenna 120 (also referred to as a “coverageantenna”) as a mobile interface to mobile device 130. In operation,donor antenna 115 is in communication with base station 125, whileserver antenna 120 is in communication with mobile devices 130.

In repeater 110, signals from base station 125 are amplified usingforward link circuitry 135, while signals from mobile device 130 areamplified using reverse link circuitry 140. Many configurations may beused for forward link circuitry 135 and reverse link circuitry 140.

There are many types of repeaters. In some repeaters, both the networkand mobile interfaces are wireless; while in others, a wired networkinterface is used. Some repeaters receive signals with a first carrierfrequency and transmit amplified signals with a second different carrierfrequency, while others receive and transmit signals using the samecarrier frequency. For “same frequency” repeaters, one particularchallenge is managing the feedback that occurs since some of thetransmitted signal can leak back to the receive circuitry and beamplified and transmitted again.

Existing repeaters manage feedback using a number of techniques; forexample, the repeater is configured to provide physical isolationbetween the two antennae, filters are used, or other techniques may beemployed.

SUMMARY

According to one embodiment of the present invention, a method forestimating a feedback channel for a wireless repeater in a wirelesscommunication system where the wireless repeater has a first antenna anda second antenna to receive a receive signal and transmit an amplifiedsignal and the receive signal is a sum of a remote signal to be repeatedand a feedback signal resulting from the feedback channel between thefirst and second antenna of the wireless repeater includes (a) usingblocks of N samples of the amplified signal as a pilot signal, (b) usingblocks of N samples of the receive signal, (c) estimating an errorcorrection term using a most recent channel estimate, (d) canceling theerror correction term from a current block of the receive signal, and(e) estimating the feedback channel between the first antenna and thesecond antenna using frequency domain channel estimation and using acurrent block of the pilot signal and the corrected block of the receivesignal, the channel estimate obtained becoming the most recent channelestimate.

According to another aspect of the present invention, a machine-readablemedium comprising instructions, which, when executed by a machine, causethe machine to perform operations, the instructions include (a) usingblocks of N samples of the amplified signal as a pilot signal, (b) usingblocks of N samples of the receive signal, (c) estimating an errorcorrection term using a most recent channel estimate, (d) canceling theerror correction term from a current block of the receive signal, and(e) estimating the feedback channel between the first antenna and thesecond antenna using frequency domain channel estimation and using acurrent block of the pilot signal and the corrected block of the receivesignal, the channel estimate obtained becoming the most recent channelestimate.

According to yet another aspect of the present invention, acomputer-readable medium including program code stored thereon includingprogram code to use blocks of N samples of the amplified signal as apilot signal, program code to use blocks of N samples of the receivesignal, program code to estimate an error correction term using a mostrecent channel estimate, program code to cancel the error correctionterm from a current block of the receive signal, and program code toestimate the feedback channel between the first antenna and the secondantenna using frequency domain channel estimation and using a currentblock of the pilot signal and the corrected block of the receive signal,the channel estimate obtained becoming the most recent channel estimate.

According to another aspect of the present invention, a wirelessrepeater having a first antenna and a second antenna to receive areceive signal and transmit an amplified signal where the receive signalis a sum of a remote signal to be repeated and a feedback signalresulting from a feedback channel between the first antenna and thesecond antenna includes a channel estimation module configured toestimate an error correction term using a most recent channel estimateand a current block of N samples of the pilot signal, to cancel theerror correction term from a current block of N samples of the receivesignal, and to estimate the feedback channel between the first antennaand the second antenna using frequency domain channel estimation andusing the current block of the pilot signal and the corrected block ofthe receive signal, the channel estimate obtained becoming the mostrecent channel estimate.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a simplified diagram of a repeater according to the prior art.

FIG. 2 shows a diagram of a repeater environment according to someembodiments of the current disclosure.

FIG. 3 is a block diagram of a repeater implementing interferencecancellation according to one embodiment of the present invention.

FIG. 4 is a flowchart illustrating the channel estimation method forcancelling multiplicative error terms which can be implemented in therepeater of FIG. 3 according to one embodiment of the present invention.

FIG. 5 is a flowchart illustrating the channel estimate method forcancelling the ICI/ISI error terms which can be implemented in therepeater of FIG. 3 according to one embodiment of the present invention.

FIG. 6 is a detail block diagram of a repeater in which the channelestimation method for sampling both causal and non-causal taps can beimplemented according to one embodiment of the present invention.

FIG. 7 illustrates the relationship between the receive signal, thepilot signal and the feedback channel.

FIG. 8 illustrates the channel response of the channel estimate withoutthe delay adjustment to left shift the pilot signal samples.

FIG. 9 illustrates the channel response of the channel estimate afterthe delay adjustment to left shift the pilot signal samples.

FIG. 10 illustrates the channel response of the channel estimate showingthe causal taps and the non-causal taps to be used by the channelestimation method according to one embodiment of the present invention.

FIG. 11 is a flowchart illustrating the channel estimation method forkeeping both the causal and the non-causal taps of the feedback channelestimate according to one embodiment of the present invention.

FIG. 12 is a flowchart illustrating the channel estimation pruningmethod in a wireless repeater according to one embodiment of the presentinvention.

FIG. 13 is a flowchart illustrating the channel estimation pruningmethod in a wireless repeater according to an alternate embodiment ofthe present invention.

DETAILED DESCRIPTION

The nature, objectives, and advantages of the disclosed method andapparatus will become more apparent to those skilled in the art afterconsidering the following detailed description in connection with theaccompanying drawings.

Prior art repeaters such as those described above may providesignificant advantages for cellular telephone or similar networks.However, existing repeater configurations may not be suitable for someapplications. For example, existing repeater configurations may not besuitable for indoor coverage applications (e.g., repeating signals for aresidence or business environment) which may be more difficult to obtainthe desired isolation between the repeater's antennas. Moreover, in sometraditional repeater implementations, the target is to achieve as high again as reasonable while maintaining a stable feedback loop (loop gainless than unity). However, increasing the repeater gain rendersisolation more difficult due to the increased signal leaking back intothe donor antenna. In general, loop stability demands require that thesignal leaking back into the donor antenna from the coverage antenna bemuch lower than the remote signal (the signal to be repeated), whetherdue to spatial isolation or other techniques such as digitalcancellation. The maximum achievable signal to interference/noise ratio(SINR) at the output of the repeater is then the same as the SINR of theremote signal at the input to the repeater. High gain and improvedisolation form two demands required for modern day repeaters, especiallythose for indoor applications.

Systems and techniques herein provide for wireless repeaters withimproved isolation between the repeaters' donor antenna (“the receivingantenna” for the example of a forward link transmission) and thecoverage antenna (“the transmitting antenna” for forward linktransmissions). Furthermore, in some embodiments, systems and techniquesherein provide for a unique repeater design employing interferencecancellation or echo cancellation to significantly improve theisolation. In some embodiments, the interference cancellation and echocancellation are realized using improved channel estimation techniquesprovided herein for accurate estimation of the channel. Effective echocancellation requires very accurate channel estimation of the leakagechannel. In general, the more accurate the channel estimate, the higherthe cancellation and hence the higher the effective isolation. Herein,“interference cancellation” or “echo cancellation” refers to techniquesthat reduce or eliminate the amount of leakage signal between repeaterantennas; that is, “interference cancellation” refers to cancellation ofan estimated leakage signal, which provides for partial or completecancellation of the actual leakage signal.

FIG. 2 shows a diagram of an operating environment 200 for a repeater210 according to embodiments of the current disclosure. The example ofFIG. 2 illustrates forward link transmissions; i.e., a remote signal 140from a base station 225 is intended for a mobile device 230. A repeater,such as repeater 210, may be used in environment 200 if an un-repeatedsignal along the path 227 between base station 225 and mobile device 230would not provide sufficient signal for effective voice and/or datacommunications received at mobile device 230. Repeater 210 with a gain Gand a delay Δ is configured to repeat a signal received from basestation 225 on a donor antenna 215 to mobile device 230 using a serverantenna 220. Repeater 210 includes forward link circuitry for amplifyingand transmitting signals received from the base station 225 to mobiledevice 230 through donor antenna 215 and server antenna 220. Repeater210 may also include reverse link circuitry for amplifying andtransmitting signals from mobile device 230 back to base station 225. Atrepeater 210, the remote signal s(t) is received as an input signal andthe remote signal s(t) is repeated as a repeated or amplified signaly(t) where y(t)=√{square root over (G)}s(t−Δ). Ideally, the gain G wouldbe large, the inherent delay Δ of the repeater would be small, the inputSNR would be maintained at the output of repeater 210 (this can be ofparticular importance for data traffic support), and only desiredcarriers would be amplified.

In practice, the gain of repeater 210 is limited by the isolationbetween donor antenna 215 and server antenna 220. If the gain is toolarge, the repeater can become unstable due to signal leakage. Signalleakage refers to the phenomenon where a portion of the signal that istransmitted from one antenna (in FIG. 2, server antenna 220) is receivedby the other antenna (in FIG. 2, donor antenna 215), as shown by thefeedback path 222 in FIG. 2. In other words, signal leakage is a resultof the transmitted signal not being totally blocked by antenna isolationbetween the receiving and transmitting antennas. Without interferencecancellation or other techniques, the repeater would amplify thisfeedback signal, also referred to as the leakage signal, as part of itsnormal operation, and the amplified feedback signal would again betransmitted by server antenna 220. The repeated transmission of theamplified feedback signal due to signal leakage and high repeater gaincan lead to repeater instability. Additionally, signal processing inrepeater 210 has an inherent non-negligible delay A. The output SINR ofthe repeater is dependent on RF non-linearities and other signalprocessing. Thus, the aforementioned ideal repeater operationalcharacteristics are often not attained. Finally, in practice, thedesired carriers can vary depending on the operating environment ormarket in which the repeater is deployed. It is not always possible toprovide a repeater that amplifies only the desired carriers.

In embodiments of the current disclosure, a repeater suitable for indoorcoverage (e.g., business, residential, or similar use) is provided. Therepeater has an active gain of about 70 dB or greater which is anexample of a sufficient gain for coverage in a moderately sizedresidence. Furthermore, the repeater has a loop gain of less than onefor stability (loop gain being referred to as the gain of the feedbackloop between the transmitting antenna and the receiving antenna) and asufficient amount of margin for stability and low output noise floor. Insome embodiments, the repeater has a total isolation of greater than 80dB. In some embodiments, the repeater employs interference/echocancellation to achieve a high level of active isolation, which issignificantly more challenging than the requirements of availablerepeaters.

Some techniques of the current disclosure utilize channel estimation toenable the required level of echo cancellation. By estimating thefeedback channel (the channel between the antennas) to a sufficientdegree of accuracy, the residual error, post echo cancellation, can besufficiently below the remote signal to realize the desired loop gainmargin for stability.

The communication system in which the repeater of the present inventioncan be deployed includes various wireless communication networks basedon infrared, radio, and/or microwave technology. Such networks caninclude, for example, a wireless wide area network (WWAN), a wirelesslocal area network (WLAN), a wireless personal area network (WPAN), andso on. The term “network” and “system” are often used interchangeably. AWWAN may be a Code Division Multiple Access (CDMA) network, a TimeDivision Multiple Access (TDMA) network, a Frequency Division MultipleAccess (FDMA) network, an Orthogonal Frequency Division Multiple Access(OFDMA) network, a Single-Carrier Frequency Division Multiple Access(SC-FDMA) network, a Long Term Evolution (LTE) network, a WiMAX (IEEE802.16) network and so on. A CDMA network may implement one or moreradio access technologies (RATs) such as cdma2000, Wideband-CDMA(W-CDMA), and so on. Cdma2000 includes IS-95, IS-2000, and IS-856standards. A TDMA network may implement Global System for MobileCommunications (GSM), Digital Advanced Mobile Phone System (D-AMPS), orsome other RAT. GSM and W-CDMA are described in documents from aconsortium named “3rd Generation Partnership Project” (3GPP). Cdma2000is described in documents from a consortium named “3rd GenerationPartnership Project 2” (3GPP2). 3GPP and 3GPP2 documents are publiclyavailable. A WLAN may be an IEEE 802.11x network, and a WPAN may be aBluetooth network, an IEEE 802.15x, or some other type of network. Thetechniques may also be implemented in conjunction with any combinationof WWAN, WLAN and/or WPAN.

Channel Estimation Techniques

Accurate channel estimation is important for high fidelity basebandcancellation of the leakage signal in on-frequency repeaters. In atypical repeater, the pilot signal used for channel estimation is theamplified signal being transmitted to the mobile device (downlink) orthe base station (uplink). The amplified signal leaks back from thetransmit antenna to the receive antenna, resulting in a feedback signalbeing added to the remote signal at the repeater input. In aninterference cancellation repeater, the feedback signal is estimated andthen cancelled out. This increases the effective isolation between thedonor and coverage antennas. If the estimate of the feedback channel issufficiently accurate, the feedback signal can be almost completelysubtracted out from the repeater input signal. The more accurate thechannel estimate, the more amplification of the output signal therepeater can sustain while maintaining the required isolation forstability. In other words, the accuracy of the repeater's channelestimate and the repeater's achievable gain are directly related.

According to one aspect of the present invention, an echo cancellationrepeater implements channel estimation in the frequency domain.Frequency domain channel estimation provides particular advantages suchas reduced complexity and increased robustness. However, frequencydomain channel estimation through the use of FFT-IFFT type processingtypically relies on a cyclic prefix in the pilot signal for maintainingorthogonality. The problem with applying frequency domain channelestimation in repeater applications is that the “pilot signal” is reallyjust the signal to be transmitted (or the amplified signal) and there isno inserted cyclic prefix in the “pilot” signal to ensure orthogonalityof the different “frequency bins”. Lack of a cyclic prefix, orequivalently having a channel that is longer that the cyclic prefix, iswell known to introduce error terms in the channel estimation, includingadditive and multiplicative errors such as inter-symbol interference(ISI) and inter-carrier interference (ICI), hence degrading the accuracyof the channel estimation.

Systems and methods of the present invention provided herein enable theuse of frequency domain channel estimation in an echo cancellationrepeater in the absence of a cyclic prefix in the pilot signal which isthe transmitted signal. Channel estimation performance degradation as aresult of the lack of a cyclic prefix in the pilot signal is mitigatedthrough the estimation and cancellation of the additive errors andmultiplicative errors. Systems and methods for improving the accuracy offrequency domain channel estimation in an echo cancellation repeaterthrough symbols pruning, sampling of non-causal taps and othertechniques are also described.

Frequency domain channel estimation in a repeater using FFT-IFFT typeprocessing on a pilot signal without cyclic prefix presents challengesin terms of additive and multiplicative noise errors, such as ICI andISI error terms. The ICI and ISI error terms arise as a result of theconvolution becoming linear rather than circular due to the lack ofcyclic prefix and because the pervious symbol leaks into the new symbolwhen there is no cyclic prefix. In general, assume h is the perfectchannel estimate of the feedback channel for a given channel tap in arepeater, the actual channel estimate for the given channel tap as aresult of the lack of cyclic prefix becomes: α*h+m+z, where m representsrandom noise, z represents the additive components of the ICI and ISIerror terms, and a represents the multiplicative component of the ICIerror term. Typically, the multiplicative error component a is veryclose to 1 but can affect accuracy in the estimation is a deviates fromthe value of 1. More specifically, the multiplicative noise errorintroduces bias to the channel estimate.

1. Cancellation of Multiplicative Errors

The use of frequency domain channel estimation in the absence of acyclic prefix in the pilot signal introduces multiplicative error thatdegrade the accuracy of the channel estimation. The multiplicative error“α” is given for each channel tap in the time domain. The multiplicativeerror is especially a problem when small block size is used for the FFTprocessing. In some embodiments of the present invention, a method toestimate and correct for the multiplicative error terms due to the useof frequency domain channel estimation in the absence of a cyclic prefixin the pilot signal (the amplified or transmit signal) is described.More specifically, the multiplicative error terms are manifested asscaling factors on the channel estimate. Accordingly, in someembodiments of the present invention, a channel estimation methodperforms scaling of a time domain channel estimate obtained from afrequency domain channel estimation process in the absence of a cyclicprefix in the pilot signal to correct the multiplicative error terms.The scaling operation applies a set of scalar multiplication factors(“scaling factors” or “correction scaling factors”) to the time domainchannel estimate obtained from the frequency domain channel estimationprocess. In some embodiments, the scaling factors depend only on the tapindex and the block size used for the frequency domain channelestimation process.

In one embodiment, the frequency domain channel estimation process usesFFTs (Fast Fourier Transforms) to transform time domain signals intofrequency domain signals, and obtains an estimate of the channel in thefrequency domain by processing these frequency domain signals. Thechannel estimate in time domain obtained as a result of taking the IFFT(Inverse Fast Fourier Transforms) of the frequency domain channelestimate is a scaled replica of the channel being estimated (alsoreferred to as the “original channel”), where each tap has a differentscaling factor. To correct for the multiplicative error terms due to thelack of a cyclic prefix in the pilot signal (the transmit signal), thetime domain channel estimate obtained from the IFFT operation is scaledusing scaling factors that are function of only the tap index and theFFT block size or block length.

The set of scalar multiplication factors (or “scaling factors”) isderived as follows. In the time domain, the channel estimation signalmodel assumes that an output signal y is the convolution of the feedbackchannel h with an input signal x, plus noise. That is, y=h*x+z, where zdenotes the noise term. The input signal x is often referred to as thepilot signal for channel estimation. In an echo cancellation repeater,the input signal x of the channel estimation signal model is the echocancelled transmit signal or the amplified signal of the repeater. Theoutput signal y of the channel estimation signal model is the receivesignal at the repeater. In other words, the receive signal y at therepeater is the pilot signal x (transmit signal) transmitted through thefeedback channel h plus noise z. The feedback channel h may include oneor more channel coefficients, associated with one or more delay paths inthe channel.

Assuming that the FFT uses a block size of N and vectorizing, thereceive signal can be expressed as y[k]=h*x[k], where k is [1 . . . N]for a FFT block length of N. The feedback channel h may include multipledelay paths. Using standard matrix arithmetic, the ICI error term can besplit up into an additive error component and a multiplicative errorcomponent. The multiplicative error component α_(n) of the ICI errorterm for a given tap n is given as:

${\alpha_{n} = {1 - \frac{n}{N}}},$where n is the tap index (0 to N−1) and N is the block size of the FFT.

In sum, when channel estimation is performed in the frequency domain inthe absence of a cyclic prefix, multiplicative error terms result andcan be estimated to be

${1 - \frac{n}{N}},$where n is the tap index and N is the FFT block size. In the channelestimation method of the present invention, the multiplicative errorterms are cancelled out by scaling the time domain channel estimateusing a set of scalar multiplication factors given as

$\left( {1 - \frac{n}{N}} \right)^{- 1}.$

In embodiments of the present invention, a wireless repeaterincorporates a channel estimation module configured to perform frequencydomain channel estimation using the channel estimation method describedabove for cancelling multiplicative errors in the channel estimate. FIG.3 is a block diagram of a repeater implementing interferencecancellation according to one embodiment of the present invention.Referring to FIG. 3, a repeater 250 receives an input signal on a donorantenna 270 for downlink communications. Repeater 250 also transmit anoutput signal on a server antenna 275 for downlink communications.Repeater 250 includes an echo canceller 252 for implementingcancellation of the feedback signal. Repeater 250 amplifies the echocancelled transmit signal at a gain stage 254 providing a variable gainG. Repeater 250 also includes a channel estimation module 260 forestimating the feedback channel 268 (denoted as “h”) between antennas275 and 270. Channel estimation module 260 provides a channel estimate ĥto echo canceller 252 to enable the estimation and cancellation of theundesired feedback signal.

In operation, repeater 250 receives the receive signal y being the sumof a remote signal to be amplified and a feedback signal (or leakagesignal) being the version of the transmitted signal that leaks back fromthe transmit antenna into the receive antenna. For the purpose ofchannel estimation, the transmit signal x, or a signal indicative of thetransmit signal x, is used as the pilot signal and the remote signal istreated as noise. Repeater 250, being an echo cancellation repeater,operates to estimate the feedback signal in order to cancel out theundesired feedback signal component in the receive signal. To that end,channel estimation module 260 generates a feedback channel estimate ĥfor echo canceller 252. Echo canceller 252 generates a feedback signalestimate based on the feedback channel estimate. Echo canceller 252subtracts the feedback signal estimate from the receive signal y togenerate the echo cancelled transmit signal x. As long as the feedbacksignal estimate is accurate, the undesired feedback signal is removedfrom the receive signal and echo cancellation is realized. In thepresent embodiment, the echo cancelled transmit signal x is coupled to avariable gain stage 254 providing a gain of G to the echo cancelledtransmit signal before transmission on antenna 275. FIG. 3 illustratesonly elements that are relevant to operation of the channel estimationmethod of the present invention. Repeater 250 may include other elementsnot shown in FIG. 3 but known in the art to realize the completerepeater operation.

The channel estimation module 260 performs frequency domain channelestimation. The results of the frequency domain channel estimation isconverted to a time domain channel estimate (such as through IFFT) thatmay include multiplicative errors due to the lack of a cyclic prefix inthe pilot signal. Channel estimation module 260 operates to scale thetime domain channel estimate obtained from the frequency domain channelestimation process using a set of scalar multiplication factors(“scaling factors” or “correction scaling factors”) where the scalarmultiplication factors are function of only the tap index and the FFTblock size, as described above. The scaled time domain channel estimateĥ can then be used by echo canceller 252 for echo cancellation. Thescalar multiplication factors have the effect of improving the accuracyof the channel estimation by canceling the multiplicative error termsresulted from the frequency domain channel estimation on a pilot signalin the absence of a cyclic prefix.

FIG. 4 is a flowchart illustrating the channel estimation method forcancelling multiplicative error terms which can be implemented in therepeater of FIG. 3 according to one embodiment of the present invention.Referring to FIG. 4, channel estimation method 300 starts by receivingthe receive signal y and the transmit signal x at the channel estimationmodule (step 302). The channel estimation module performs frequencydomain channel estimation (step 304). In one embodiment, the channelestimation module performs Fast Fourier Transform (FFT) on N-sampleblocks of the transmit signal as the pilot signal and N-sample blocks ofthe receive signal. As a result of the FFT operation, frequency domainpilot signal blocks and frequency domain receive signal blocks aregenerated from blocks of N samples of the pilot signal and the receivesignal. A channel estimate in the frequency domain is obtained fromthese frequency domain samples. Then, the channel estimation methodobtains a time domain channel estimate from the frequency domain channelestimate, such as by performing Inverse Fast Fourier Transform (IFFT)operations (Step 306). The IFFT is performed on a signal generated byone or more signal processing operations on the frequency domain pilotblocks and the frequency domain receive signal blocks. Finally, thechannel estimation method scales the time domain channel estimate usinga set of correction scaling factors (step 308). In one embodiment, thecorrection scaling factors are function of only the tap index and theFFT block size. The resulting scaled channel estimate can then be usedfor echo cancellation in the repeater.

In one embodiment, the scaling of the time domain channel estimate isapplied after the IFFT operation. In other embodiments, the scaling ofthe time domain channel estimate can also be carried out afteraveraging, or other signal processing operations such as truncation orthresholding operations.

In repeater 250, channel estimation module 260 may be implemented insoftware, hardware, firmware, or some combination, and may be configuredto perform the functions described in the embodiments described herein.In a software implementation, channel estimation module 260 may comprisememory and one or more processors that may be dedicated to the channelestimation module 260 or be shared with other features or modules of therepeater. N-sample blocks indicative of the transmit signal may bestored in memory, as well as N-sample blocks indicative of the receivesignal. The processor may access instructions to read the stored dataand perform FFT operations on the data to generate frequency domainpilot signal blocks and frequency domain receive signal blocks. Theprocessor may access instructions to generate a frequency domain channelestimate from the frequency domain pilot and receive signal blocks. Theprocessor may access instructions to perform an inverse fast Fouriertransform on the frequency domain channel estimate to generate a timedomain channel estimate. In some embodiments, the processor may accessinformation indicative of one or more scaling factors to scale the timedomain channel estimate, while in some embodiments scaling is performedprior to the IFFT operations. Similarly, in a hardware implementation,the FFT and IFFT operations may be performed using hardware such as adigital signal processor (DSP).

By scaling the time domain channel estimate using the scalarmultiplication factors described above, the multiplicative error term inthe frequency domain channel estimation is eliminated and the channelestimation performance is improved.

2. Cancellation of ICI/ISI Error Terms

As described above, the use of frequency domain channel estimation inthe absence of a cyclic prefix introduces errors such as inter-carrierinterference (ICI) and inter-symbol interference (ISI) that degrade theaccuracy of the channel estimation. More specifically, the ICI/ISIerrors arise as a result of the difference between a linear convolutionand a circular convolution in the frequency domain channel estimationprocess due to the lack of a cyclic prefix in the pilot signal. If thereis no cyclic prefix, the previous symbol will leak into the current one,causing inter-symbol interference. According to embodiments of thepresent invention, a method to estimate the ICI and ISI error terms inthe frequency domain channel estimate and remove the error terms fromthe channel estimate is provided to improve the accuracy of the channelestimation. The channel estimation method for canceling ICI/ISI errorscan be implemented in an echo cancellation repeater such as repeater 250of FIG. 3. For instance, the channel estimation module 260 can implementthe channel estimation method of the present invention for cancelingICI/ISI errors due to the use of frequency domain channel estimation inthe absence of a cyclic prefix in the pilot signal.

In frequency domain channel estimation processing, symbols of the pilotsignal (the transmit signal), as well as symbols of the receive signal,are grouped in blocks of length N, where N is the size of the FFT beingperformed on both the pilot symbols and the receive symbols. The receivesymbols are assumed to be the linear convolution of the pilot symbolsand the channel, plus noise. Because there is no cyclic prefix, thepilot symbols of the previous block leak into the current receivesymbols block, causing ISI (inter-symbol interference). Similarly, ICI(inter-carrier-interference) is also created on account of the lack of acyclic prefix. In the absence of a cyclic prefix, the receive symbolsare just a linear convolution of the pilot symbols and the channel. Ifthere was a cyclic prefix in the pilot symbols, the received symbolswould be the circular convolution of the current block of pilot symbolsand the channel. The difference between circular and linear convolutionresults in ICI error terms.

More specifically, let H(f) denotes the perfect feedback channelestimate, P(f) denotes the Fast Fourier Transform of one block of Nsamples of the pilot signal, that is, P(f)=FFT(pilot), and R(f) denotesthe Fast Fourier Transform of one block of N samples of the receivesignal, that is, R(f)=FFT(rx_signal), the feedback channel estimate Ĥ(f)for a particular frequency, if computed only using one block of pilotsignal and one block of receive signal, is given as:

$\begin{matrix}{{\hat{H}(f)} = \frac{{P(f)}*{R(f)}}{{P(f)}*{P(f)}}} \\{= \frac{{P(f)}*\left( {{feedbacksignal} + {remotesignal}} \right)}{{P(f)}*{P(f)}}} \\{= {\frac{{P(f)}*\left( {{{H(f)}{P(f)}} + {remotesignal}} \right)}{{P(f)}*{P(f)}} + {I\; C\; I} + {I\; S\; I}}} \\{{= {{H(f)} + \frac{{P(f)}*{remotesignal}}{{P(f)}*{P(f)}} + {I\; C\; I} + {I\; S\; I}}},}\end{matrix}$where “P(f)*R(f)” denotes conjugate of the pilot symbols and the receivesymbols at the particular frequency. The use of frequency domain channelestimation in the absence of a cyclic prefix introduces the error termsICI and ISI as shown above. These error terms degrade the accuracy ofthe channel estimation.

In accordance with embodiments of the present invention, the ICI errorterms are reconstructed using the most recent channel estimate, andintroduced to the current receive symbol block so that the receivesymbols block approximately equals to the circular convolution of thecurrent pilot symbols block and the feedback channel. Similarly, the ISIterms are reconstructed using the most recent channel estimate andintroduced to the current receive symbols block to minimize the leakagefrom the previous block of pilot symbols. The removal of the ISI termsalso renders the receive symbols block approximately equals to thecircular convolution of the current pilot symbols block and the feedbackchannel. In this manner, ICI and ISI error terms in the channel estimateare cancelled out and a more accurate channel estimate is obtained.

In one embodiment, the ICI error term and the ISI error term arereconstructed as follows. Using the same channel estimation signal modeldescribed above, the receive signal y can be modeled as a linearconvolution of the transmit signal x (the pilot signal) with thefeedback channel h plus noise. More specifically, assuming the feedbackchannel has a length of L and is given as h=[h(0) h(1) . . . h(L)], thereceive signal sample y(n) can be expressed as:

${{y(n)} = {{\sum\limits_{k = 0}^{L}{{x\left( {n - k} \right)}{h(k)}}} + {z(n)}}},$where z(n) denotes the noise samples. The channel estimation processestimates the channel h using the receive samples y(n) assuming that thetransmit (pilot) samples x(n) are known.

In frequency domain channel estimation, such as FFT-IFFT typeprocessing, the receive samples y(n) are divided into blocks of length Nand the i-th block of the receive samples is expressed as:y _(N) ^(i) =y ^(i) =[y ₁ ^(i) ,y ₂ ^(i) Ly _(N) ^(i)]^(T).

A linear convolution matrix L^(i) for the i-th block of the pilotsamples x(n) is given as:

$L_{N \times N}^{i} = {L^{i} = {\begin{pmatrix}x_{1}^{i} & x_{N}^{i - 1} & x_{N - 1}^{i - 1} & L & x_{1}^{i - 1} \\x_{2}^{i} & x_{1}^{i} & x_{N}^{i - 1} & L & x_{2}^{i - 1} \\x_{3}^{i} & x_{2}^{i} & x_{1}^{i} & L & x_{3}^{i - 1} \\M & M & M & O & M \\x_{N}^{i} & x_{N - 1}^{i} & x_{N - 2}^{i} & L & x_{1}^{i}\end{pmatrix}.}}$

A circular convolution matrix C^(i) for the i-th block of the pilotsamples x(n) is given as:

$C_{N \times N}^{i} = {C^{i} = {\begin{pmatrix}x_{1}^{i} & x_{N}^{i} & x_{N - 1}^{i} & L & x_{1}^{i} \\x_{2}^{i} & x_{1}^{i} & x_{N}^{i} & L & x_{2}^{i} \\x_{3}^{i} & x_{2}^{i} & x_{1}^{i} & L & x_{3}^{i} \\M & M & M & O & M \\x_{N}^{i} & x_{N - 1}^{i} & x_{N - 2}^{i} & L & x_{1}^{i}\end{pmatrix}.}}$

Due to the lack of cyclic prefix in the pilot samples x(n), theconvolution with the feedback channel becomes a linear convolutionrather than the ideal circular convolution. The receive sample blocky^(i) can be written in matrix-vector notation using the linearconvolution matrix L^(i) as follows:y ^(i) =L ^(i) h+z ^(i).

Because the receive samples y(n) should ideally be a circularconvolution of the pilot samples with the feedback channel, not a linearconvolution, the receive sample block y^(i) can be expressed using thecircular convolution matrix and the linear convolution matrix asfollows:y ^(i) =C ^(i) h+(L ^(i) −C ^(i))h+z ^(i).

The first term in the above expression is the ideal circular convolutionof the pilot samples with the feedback signal. The second term in theabove expression represents the ICI/ISI error terms resulted from thedifference between the circular convolution and linear convolution.Since both the linear convolution matrix L^(i) and the circularconvolution matrix C^(i) are based on the pilot samples and are known,the ICI and ISI error terms can be computed using the latest channelestimate h. The computed error terms can then be subtracted from thereceive sample block to remove the ICI/ISI errors and the channelestimate can be computed using the corrected receive sample block toyield a more accurate channel estimate.

In one embodiment, the ICI and ISI error terms are eliminated bysubtracting the circular convolution matrix from the linear convolutionmatrix first (L^(i)−C^(i)) multiplying by the latest channel estimate ĥand then subtracting the difference (L^(i)−C^(i))*ĥ from the receivesample block y^(i). In this case, the error is subtracted in the timedomain. The channel would then be estimated using frequency domainprocessing.

Applying frequency domain processing, such as by taking the DiscreteFourier Transform (DFT) of the receive sample block y^(i) using an N×NDFT matrix F, the receive sample block in frequency domain is given as:Y ^(i)=diag(X ^(i))H+F(L ^(i) −C ^(i))h+Z ^(i),where X^(i), H and Z^(i) are the DFT of x^(i), h and z^(i),respectively.

To eliminate the ICI and ISI error terms, the ICI and ISI error termsare subtracted from the receive sample block. The subtraction can takeplace in the time domain or in frequency domain. Assuming {tilde over(h)} is the latest channel estimate or the most recent channel estimate,the latest channel estimate can be used to calculate an updated channelestimate as follows:Y ^(i) −F(L ^(i) −C ^(i)){tilde over (h)}=diag(X ^(i))H+F(L ^(i) −C^(i))(h−{tilde over (h)})+Z ^(i).By incorporating the F(L^(i)−C^(i)){tilde over (h)} error correctionterm in the receive sample block, the error in the channel estimate iscancelled. The result of introducing the error correction term in thereceive sample block is to make the receive sample block look more likethe result of a circular convolution. In this example, the error term issubtracted in the frequency domain.

To compute the channel estimate, if only using one block of pilotsamples and one block of receive samples, the frequency domain channelestimate Ĥ is given as:{circumflex over (H)}=diag(X ^(i))⁻¹ Y _(C) ^(i),where Y_(C) ^(i) denotes the corrected Y^(i) being(Y^(i)−F(L^(i)−C^(i)){tilde over (h)}). In this case, the error term onthe channel estimation can be expressed as:diag(X^(i))⁻¹*F(L^(i)−C^(i))*{tilde over (h)}. This error term on thechannel estimate can be subtracted out directly from the channelestimate. According to another aspect of the present invention, thecomputed error correction term is processed through the channelestimation process and then the processed channel estimate error term issubtracted from the computed channel estimate.

The time domain channel estimate is formed by taking the inverse DFT ofĤ and is given as:ĥF ⁻¹ Ĥ=[ĥ ₀ ,ĥ ₁ ,ĥ ₂ ,Lĥ _(M) ,ĥ _(M) ,ĥ _(M+1) ,ĥ _(N−1)].The larger the FFT block size N, and the smaller the channel length L,the better the channel estimate Ĥ is.

In the above description, the frequency domain channel estimationprocess is performed using a Discrete Fourier Transform (DFT). In otherembodiments, the frequency domain channel estimate process can beperformed using a Fast Fourier Transform (FFT). The use of a specifictype of frequency domain signal processing is not critical to thepractice of the present invention.

FIG. 5 is a flowchart illustrating the channel estimate method forcancelling the ICI/ISI error terms which can be implemented in therepeater of FIG. 3 according to one embodiment of the present invention.Referring to FIG. 5, channel estimation method 350 starts by receivingblocks of N samples of the transmit signal (the pilot signal) and thereceive signal (step 352). Then, using the block of N pilot samples anda current channel estimate or the most recent channel estimate, method350 estimates an error correction term. In the present embodiment,method 350 computes a linear convolution matrix L^(i) (step 354) andalso computes a circular convolution matrix C (step 356). Then, theerror correction term is computed using the linear and circularconvolution matrices and the most recent channel estimate (step 358).More specifically, the error correction term is (L^(i)−C^(i)){tilde over(h)} in the time domain and F(L^(i)−C^(i)){tilde over (h)} in thefrequency domain where h denotes the latest or most recent channelestimate and F is a frequency domain transform. Then, the errorcorrection term is subtracted from the receive sample block (step 360).The subtraction can be carried out in the time domain or in thefrequency domain. An updated channel estimate is then computed using thecorrected receive sample block and the pilot sample block in frequencydomain (step 362). In some embodiments, the updated channel estimate iscomputed using the current block of corrected receive samples and pilotsamples as well as previous blocks of similarly corrected receivesamples and their corresponding pilot samples. The most recent channelestimate is then replaced with the updated channel estimate (step 364).

In an alternate embodiment, the channel estimation method for ICI/ISIerror terms reconstruction and removal is repeated on the next block ofN samples of the transmit signal and the receive signal using the latestor most recent channel estimate to compute the error correction term toimprove the channel estimate (step 366). In each repetition, the ICI/ISIerror correction term is computed using the latest channel estimate andthen removed from the receive samples to provide an improved channelestimate. In another alternate embodiment, the channel estimation methodfor ICI/ISI error terms reconstruction and removal is iterated on thecurrent block of N samples of the transmit signal and the receive signalfor a fixed number of iterations (step 368). In each iteration, theICI/ISI error correction term is computed using the latest channelestimate and then removed from the receive samples to provide animproved channel estimate. Multiple iterations of the method of thepresent invention enhance the accuracy of the channel estimate. Then,after the current sample blocks have been iterated for the fixed numberof iterations, the channel estimation method repeats on the next blockof N samples of the transmit signal and the receive signal using thelatest or most recent channel estimate to compute the error correctionterm. In FIG. 5, step 368 illustrates an alternate embodiment of thechannel estimation which is optional and may be omitted if iteration onthe same sample blocks is not desired.

As described above, channel estimation module 260 in repeater 250 may beused to implement the channel estimate method of the present inventionfor cancelling the ICI/ISI error terms. Channel estimation module 260may be implemented in software, hardware, firmware, or some combination,and may be configured to perform the functions described in theembodiments described herein. In a software implementation, channelestimation module 260 may comprise memory and one or more processorsthat may be dedicated to the channel estimation module 260 or be sharedwith other features or modules of the repeater. N-sample blocksindicative of the transmit signal may be stored in memory, as well asN-sample blocks indicative of the receive signal. The processor mayaccess instructions to read the stored data and compute the linearconvolution matrix and compute the circular convolution matrix. Theprocessor may further access instructions to compute the errorcorrection term and to subtract the error correction term from thereceive samples. The processor may access instructions to generateupdated channel estimates from the corrected receive sample block and toreplace the most recent channel estimate with the updated channelestimate. In some embodiments, the processor may access instructions toiterate the computation using the current receive samples and pilotsamples for a fixed number of iterations. In some other embodiments, theprocessor may access instructions to repeat the computation using thenext block of receive and pilot samples. Similarly, in a hardwareimplementation, the convolution computation and channel estimatecomputation operations may be performed using hardware such as a digitalsignal processor (DSP).

Reconstructing the ICI/ISI error terms with the latest channel estimate,and then removing these error terms from the actual channel estimatereduce the error floor in the frequency domain channel estimation causedby not having a cyclic prefix in the pilot signal. The correction ofthese error terms to the channel estimate enables the use of frequencydomain channel estimate in a wireless repeater. As described above,performing channel estimation in the frequency domain, as opposed to thetime domain, is desirable because of complexity considerations andbecause time domain channel estimation algorithms can have issues withrobustness. However, channel estimation in the frequency domain requiresa cyclic prefix in the pilot signal for accurate channel estimation. Insome applications, it is not possible to add a cyclic prefix to thepilot signal. The channel estimation method of the present inventionprovides particular advantages by mitigating the channel estimationperformance loss due to the lack of a cyclic prefix. Improvement in thechannel estimation, in the context of a repeater, allows a significantimprovement in the amount of gain achievable. Furthermore, even for agiven, fixed gain in the repeater, when the channel estimate improves,the output SNR (signal-to-noise ratio) increases (where output SNR is ameasure of the noise introduced by the repeater), meaning that thefidelity of the repeater improves.

3. Retain Non-Causal Tap

The use of frequency domain channel estimation in the absence of acyclic prefix in the pilot signal introduces ICI and ISI error termsthat degrade the accuracy of the channel estimation. More specifically,the ICI/ISI error terms arise as a result of the difference between thelinear convolution and the circular convolution due to the lack of acyclic prefix in the pilot signal.

According to embodiments of the present invention, a channel estimationmethod operates to capture both the causal and the non-causal taps ofthe channel in generating the channel estimate. By capturing thenon-causal taps as well as the causal taps of the channel, a moreaccurate channel estimate can be obtained. In some embodiments of thepresent invention, a channel estimation method for mitigating thechannel estimation performance loss due to the lack of a cyclic prefixin the pilot signal shifts the largest channel tap in the channelestimate to the reference time “0” for channel estimation. However,shifting the channel tap in this manner causes all the channel taps thatoccur before the largest tap to become “non-causal” taps. That is, partof the feedback channel to be estimated becomes non-causal. Discardingthese non-causal taps from the channel estimation can introduce apotentially large error floor which could result in instability in thefeedback loop of the repeater. Accordingly, the channel estimationmethod of the present invention operates to capture both the causal andthe non-causal taps of the channel in generating the channel estimate.

For instance, in one exemplary repeater system, discarding thenon-causal part of the channel may result in instability well before therepeater has reached full gain. However, when the non-causal taps of thechannel is sampled according to the channel estimation method of thepresent invention, a ˜20 dB output SNR gain can be achieved when therepeater reaches full gain.

FIG. 6 is a detail block diagram of a repeater in which the channelestimation method for sampling both causal and non-causal taps can beimplemented according to one embodiment of the present invention.Referring to FIG. 6, a repeater 400 receives a remote signal S(t) on adonor antenna 415 which is coupled to receive circuitry including atransceiver front end circuit 416 and a receive filter 443. The receivesignal samples r[k] from the transceiver front end circuit 416 arecoupled to receive filter (Rx filter) 443 and then to an echo cancellerincluding a summer 444 for echo cancellation. The echo cancelled receivesignal r′[k] is coupled to a delay element 446 to introduce a desiredamount of delay D1 to decorrelate the transmit signal from the remotesignal. Delay element 446 can be a fixed delay or a variable delayelement as shown in FIG. 6. In other embodiments, delay element 446 canbe provided before the echo canceller. The delayed echo cancelled signalis coupled to transmit circuitry including a transmit filter (Tx filter)448, a gain stage 449 applying a gain of G and a transceiver front endcircuit 418. Gain stage 449 generate the transmit signal samples y[k]which are provided to transceiver front end circuit 418. Repeater 400transmits the transmit signal Y(t) generated by transceiver front endcircuit 418 on a coverage antenna 420.

In repeater 400, a gain control block 480 controls the variable gain ofgain stage 449 and a channel estimation block 450 performs channelestimation of the feedback channel. The transmit signal samples y[k] areused as the pilot signal for the gain control block 480 and the channelestimation block 450. In accordance with the present invention, thechannel estimation block 450 implements frequency domain channelestimation, such as FFT-IFFT type processing or DFT-IDFT typeprocessing. Channel estimation block 450 also receives the receivesignal samples r[k] and perform channel estimation to generate afeedback channel estimate ĥ. More specifically, channel estimation block450 generates the feedback channel estimate ĥ using at least N samplesof the pilot signal and N samples of the receive signal r[k]. Thefeedback channel estimate ĥ is provided to a feedback signal estimationblock 452 which, together with the transmit signal y[k], computes afeedback signal estimate {circumflex over (l)}[k]. In the case where theecho canceller is applied to the receive signal after filtering, thefeedback signal estimate computation should also go through the samefiltering to generate the corresponding feedback signal estimate. In thepresent embodiment, the canceller 444 is positioned after the receivefilter, therefore, the feedback signal estimate {circumflex over (l)}[k]is computed as the convolution of the feedback channel estimate with thetransmit signal y[k] and also with the receive filter (Rx filter 443).The feedback signal estimate l[k] is provided to summer 444 to besubtracted from the receive signal r[k] for realizing echo cancellation.

According to the channel estimation method of the present invention, thepilot samples y[k] are shifted so that the largest channel tap is movedto a reference time “0”. To that end, a variable delay element 454introduces a delay D2 to the pilot samples before the pilot samples areprovided to the channel estimation block 450. The adjustable delay D2operates to delay the pilot signal in time and the desired amount ofdelay D2 is introduced so that the largest channel tap is aligned withthe reference time 0.

In frequency domain channel estimation processing, samples of the pilotsignal (the transmit signal), as well as samples of the receive signal,are grouped in blocks of length N, where N is the size of the frequencytransform (such as FFT). FIG. 7 illustrates the relationship between thereceive signal, the pilot signal and the feedback channel. Morespecifically, the receive signal is the pilot signal convolve with thefeedback channel h. According to embodiments of the present invention,the channel estimation method applies a variable delay, such as throughvariable delay element 454, to each block

¹,

²,

³ of N pilot samples. Thus, each block of pilot samples is delayed bydelay D2 so that:

¹=[p_(1+D2),p_(2+D2)Lp_(N+D2)]. The delayed pilot samples are then sentwith the receive samples to the channel estimation block to estimate thefeedback channel. The receive samples

¹,

²,

³ are not delayed so that:

^(i)=[r₁ ^(i),r₂ ^(i)Lr_(N) ^(i)].

As described above, the left shifting of the pilot signal causes part ofthe channel to be estimated to become non-causal. FIG. 8 illustrates thechannel response of the channel estimate without the delay adjustment toleft shift the pilot signal samples. FIG. 9 illustrates the channelresponse of the channel estimate after the delay adjustment to leftshift the pilot signal samples. Referring to FIG. 8, without any delayadjustment, the largest channel tap does not align with reference time0. However, referring to FIG. 9, after applying an appropriate amount ofdelay D2, the largest channel tap of the channel response is now shiftedto the left to be aligned with reference time 0.

FIG. 10 illustrates the channel response of the channel estimate showingthe causal taps and the non-causal taps to be used by the channelestimation method according to one embodiment of the present invention.Referring to FIG. 10, as a result of the left shifting of the pilotsignal samples, the largest channel tap is now at reference time 0.However, there is some signal energy present at the end of the FFTblock, as shown by the enlarged graph in FIG. 10. The signal energy atthe end of the FFT block is indicative of the channel taps that becomenon-causal due to the left shifting of the pilot samples.

According to embodiments of the present invention, the channelestimation method retain both the causal taps and the non-causal taps inthe time domain feedback channel estimate. In one embodiment, thechannel estimate is of a length N. For channel estimation purpose, M1causal taps and M2 non-causal taps are retained in the channel estimate.The causal taps are taken from sample 0 to M1, where M1 is assumed to bemuch less than N since the delay spread of the feedback channel for therepeater is typically small. The non-causal taps are taken from sampleN−M2 to N, where M2 is assumed to be much less than N for the samereason as M1. In one embodiment, the total channel estimate (both thecausal and the non-causal taps M1+M2) is less than N taps and only M1+M2taps of the channel estimate are kept.

Returning to FIG. 6, the receive filter Rx filter 443 lies after thereceived samples are tapped for channel estimation and before the summer444 of the echo canceller. Because the actual echo cancellation happensafter the receive filter 443, for that reason, in one embodiment, thepilot signal is convolved with the total channel estimate ĥ and with afilter, such as the receive filter, to generate the feedback signalestimate {circumflex over (l)}[k] for echo cancellation. That is, thetotal feedback channel estimate h is convolved with the receive filterto generate a “composite channel estimate” given as ĥ*RxFilter. Thecomposite channel estimate is then convolved with the pilot samples y[k]to generate the feedback signal estimate. The feedback signal estimatethus generated is then supplied to summer 444 to be subtracted from thereceive signal r[k] to realize echo cancellation.

Accordingly, in some embodiments, such as when the echo cancellation isperformed after receive signal filtering, the pilot signal needs to beconvolved with both the channel estimate as well as another filterrepresentative of the receive signal filtering, such as the receivefilter. This can be done by convolving the pilot signal with a compositechannel estimate comprised of the channel estimate as well as the otherfilter. If the other filter has a delay of D which is greater than M2,the number of non causal taps, and the other filter is circularlyconvolved with the channel estimate, then after keeping the first M1+L−1taps where L is the length of the filter, both the non-causal and causaltaps are included in the composite channel estimate.

FIG. 11 is a flowchart illustrating the channel estimation method forkeeping both the causal and the non-causal taps of the feedback channelestimate according to one embodiment of the present invention. Referringto FIG. 11, channel estimation method 500 starts by receiving blocks ofN samples of the amplified signal as a pilot signal and receiving blocksof N samples of the receive signal (step 502). Then, method 500introduces a delay to the samples of the pilot signal to align thelargest channel tap of the feedback channel to a first reference time(step 504). Method 500 then estimates the feedback channel between thefirst antenna and the second antenna using frequency domain channelestimation using samples of the pilot signal and samples of the receivesignal (step 506). Method 500 generates a time domain feedback channelestimate from the frequency domain channel estimation. In oneembodiment, the time domain feedback channel estimate having N taps(step 508). Finally, method 500 retains M1 causal taps and M2 non-causaltaps in the time domain feedback channel estimate (step 510). The M1causal taps are the first M1 taps of the channel estimate and the M2non-causal taps are the last M2 taps of the channel estimate. In oneembodiment, M1+M2 is less than N and only M1+M2 taps of the channelestimate are kept. In other embodiments of the channel estimation methodof the present invention, no delay is introduced to the pilot samplesand the channel estimation method operates in the same manner to captureboth the causal taps and the non-causal taps of the channel estimate. Inother words, delay D2 in repeater 400 may be set to zero and step 504 inmethod 500 is optional.

Channel estimation module 260 in repeater 250 may be used to implementthe channel estimate method of the present invention for keeping boththe causal and the non-causal taps of the feedback channel estimate.Channel estimation module 260 may be implemented in software, hardware,firmware, or some combination, and may be configured to perform thefunctions described in the embodiments described herein. In a softwareimplementation, channel estimation module 260 may comprise memory andone or more processors that may be dedicated to the channel estimationmodule 260 or be shared with other features or modules of the repeater.N-sample blocks indicative of the transmit signal may be stored inmemory, as well as N-sample blocks indicative of the receive signal. Theprocessor may access instructions to read the stored data and introducethe necessary delay. The processor may access instructions to performFFT operations on the data to generate frequency domain pilot signalblocks and frequency domain receive signal blocks. The processor mayaccess instructions to generate a frequency domain channel estimate fromthe frequency domain pilot and receive signal blocks. The processor mayaccess instructions to perform an inverse fast Fourier transform on thefrequency domain channel estimate to generate a time domain channelestimate. In some embodiments, the processor may access instructions toretain causal taps and non-causal taps in the time domain channelestimate. Similarly, in a hardware implementation, the FFT and IFFToperations may be performed using hardware such as a digital signalprocessor (DSP).

In alternate embodiments of the present invention, the M1 causal tapsand the M2 non-causal taps of the time domain channel estimate arescaled using the set of correction scaling factors described above withreference to FIG. 4 for cancelling multiplicative error in the channelestimate. In one embodiment, the correction scaling factors are functionof only the tap index and the FFT block size. The resulting scaledchannel estimate can then be used for echo cancellation in the repeater.

4. Truncating the Channel Estimate

When frequency domain channel estimation using FFT/IFFT processing isapplied in a repeater, the feedback channel estimate is equal to the FFTsize N, for example 1024. However, the feedback channel duration islimited. For instance, in repeater applications, the feedback channel isusually on the order of tens of samples, such as 64 samples. Channelestimation algorithm based on pilots introduces noise in the channelestimate. The noise is strong where the pilot is weak, that is,out-of-band. The noise also permeates to all samples of the time-domainchannel estimation. In a conventional repeater, the receive filter(RxFilter) is sometimes used to reduce channel estimation noise bysuppressing the noise at a stopband.

According to embodiments of the present invention, a channel estimationmethod in a repeater applies truncation to the time-domain channelestimate to zero out channel taps that do not contain the feedbackchannel so as to reduce channel estimation noise. More specifically, themethod uses a priori information about the channel length to determinein which taps the channel lies. Channel taps that are outside of thechannel are zeroed out to reduce noise.

In one embodiment, for a channel estimate of size N and a feedbackchannel having a channel length L where L is assumed to be much lessthan N, the channel estimate is truncated by zeroing out the channeltaps that are outside of the channel. For instance, in one embodiment,for a channel estimate ĥ having a size N, the truncation is applied tokeep the first M taps of the channel and to zero out channel taps M+1 toN. The truncated channel estimate is thus given as:ĥ=[ĥ ₀ ,ĥ ₁ ,ĥ ₂ ,Lĥ _(M),0L0],where M is greater than or equal to the channel length L. In thismanner, the channel estimate that lies in the first M taps of thechannel are retained while the channel estimate that lies in the M+1 toN taps of the channel estimate are zeroed out to remove noise.

In another embodiment, channel estimate truncation is applied topreserve both the causal and non-causal taps of the channel estimate. Inthe case where the pilot samples are left shifted to align the largestchannel tap at reference time 0, part of the channel becomes non-causaland the non-causal channel needs to be accounted for in the channelestimation process. In the channel estimation method described above,both the causal and the non-causal are used for the purpose of channelestimation. When channel estimate truncation is applied, the causal andnon-causal taps of the channel estimate are preserved while the rest ofthe channel estimate is zeroed out.

In one embodiment, for a channel estimate of size N, a feedback channelhaving a causal channel of length L and a non-causal channel of lengthP, where L and P are both assumed to be much less than N, the channelestimate is truncated by zeroing out the channel taps that are outsideof the causal and the non-causal channels. For instance, the truncationis applied to keep the first M1 taps and the N−M2 to N taps of thechannel while zeroing out channel taps M1+1 to N−M2−1. The truncatedchannel estimate is thus given as:ĥ=[ĥ ₀ ,ĥ ₁ ,ĥ ₂ ,Lĥ _(M1),0L0,ĥ_(N−M2) ,ĥ _(N−M2+1) ,Lĥ _(N−1)],where M1 is greater than or equal to the causal channel length L and M2is greater than or equal to the non-causal channel length P. In thismanner, the channel estimate that lies in the M1+1 to N−M2−1 taps of thechannel estimate are zeroed out to remove noise.

By truncating the part of the time domain channel estimate, channelestimate noise is reduced to improve the accuracy of the channelestimation.

5. Channel Estimate Pruning

During the operation of a same frequency repeater, it is desirable tomaintain stability in the repeater in the presence of large signaldynamics. In order to maintain repeater stability in the presence oflarge signal dynamics in the remote signal energy, the repeater needs toknow when a sudden change in receive power occurs. Sudden changes in thereceive power can corrupt the channel estimation so that the feedbackchannel estimate becomes inaccurate and echo cancellation of thefeedback signal becomes incorrect, leading to repeater instability.

According to embodiments of the present invention, a method forimproving the stability of a repeater in the presence of large signaldynamics implements channel estimation pruning. In some embodiments, thepower of the remote signal is monitored to detect sudden changes in thereceive power. The channel estimation pruning method operates to discardchannel estimates or pilot and receives samples associated with largechange in the signal dynamics of the remote signal. In this manner, theovercall channel estimate will not be corrupted due to sudden changes inthe receive power.

FIG. 12 is a flowchart illustrating the channel estimation pruningmethod in a wireless repeater according to one embodiment of the presentinvention. The channel estimation pruning method of the presentinvention can be implemented in an echo cancellation repeater, such asthe repeater shown in FIG. 3 or FIG. 6. In practice, the channelestimation pruning method is implemented in a channel estimate module inthe channel estimation block (260 or 450) of the repeaters in FIG. 3 andFIG. 6. More specifically, the channel estimation block estimates thefeedback channel using blocks of incoming pilot samples (x in FIG. 3; ordelayed y[k] in FIG. 6) and blocks of incoming receive samples (y inFIG. 3; or delayed r[k] in FIG. 6). In some embodiments, a final channelestimate is computed using the blocks of incoming pilot samples and theblocks of incoming receive samples. The final channel estimate is thenused by the echo canceller to cancel out the feedback signal. In someother embodiments, a channel estimate is computed using a block ofincoming pilot samples and a block of incoming receive samples andmultiple channel estimates thus computed are then averaged to generatethe final channel estimate. In the present description, the channelestimate computed for each individual pilot sample block and thecorresponding receive sample block is referred to as a “sub channelestimate”.

Referring to FIG. 12, a channel estimation pruning method 600implemented in a channel estimation algorithm operates to discardsamples or blocks of N samples of the pilot signal and the receivesignal used for the channel estimation after a large jump in remotesignal power has been detected, N being the size of the FFT used forfrequency domain channel estimation or the size of the feedback channelestimate. Method 600 starts by receiving the incoming receive signal ata repeater which includes the remote signal (step 602). As describedabove, the receive signal is the sum of the remote signal and thefeedback signal. A signal in the feedback loop of the repeater is usedas the pilot signal for channel estimation. In the present embodiment,the pilot signal is the echo cancelled transmit signal or the amplifiedsignal of the repeater. The pilot signal and the receive signal, orblocks of pilot signal samples and blocks of receive signal samples, areprovided to the channel estimation block for performing channelestimation (step 603).

Then, swings in the power level of the remote signal are detected todetermine if there are large power swings in the remote signal (step604). The swings in the power level of the remote signal can be detecteddirectly from the receive signal or it can be detected indirectlythrough measurements of other signals having a power level responsecorresponding to the remote signal. In one embodiment, swings in theremote signal is detected by using an FIR (finite impulse response) orIIR (infinite impulse response) filter. The detection is implemented bymonitoring the power differential between samples of the receive signalspaced a given time apart to determine if a sudden change in the powerlevel has occurred.

When no large power swings are detected in the gain control input signal(step 606), method 600 returns to step 602 to continue to receive theincoming receive signal and continue to estimate the feedback channelbased on the incoming samples of the pilot signal and the receivesignal. However, when a large power swing, such as a power swing of 9 dBor greater, is detected (step 606), samples or blocks of N samples ofthe pilot signal and the corresponding receive signal are discarded(step 608). A final channel estimate is computed using one or more ofthe undiscarded blocks of the pilot and receive signals (step 610).Method 600 then returns to step 602 to continue to receive incomingreceive signal. In this manner, samples of or blocks of N samples of thepilot signal and receive signal associated with swings in the remotesignal power are discarded before the pilot signal samples and receivesignal samples corrupt the channel estimates.

In one embodiment, method 600 discards both the current block of Nsamples and one or more previous blocks of N samples of the pilot andreceive signals in response to a large power swing being detected in theremote signal. The reason to discard both the current and the previoussamples of the pilot and receive signals is that there is always somedelay between the time a change in remote signal power occurs, and thetime the repeater can detect that change. Therefore, it is possible thatsome of the pilot signal samples and the receive signal samples arealready corrupted by the power swing. Accordingly, in channel estimationpruning method 600, both the current and the previous pilot and receivesignal samples are discarded when a large power swing is detected sothat no corrupted samples are retained and used for computed the finalchannel estimate.

In another embodiment, swings in the remote signal can be detected usingthe receive signal, or the cancelled signal at any point along the mainpath through the repeater.

FIG. 13 is a flowchart illustrating the channel estimation pruningmethod in a wireless repeater according to an alternate embodiment ofthe present invention. Referring to FIG. 13, in channel estimationpruning method 700, the channel estimation pruning method operates todiscard sub channel estimates after a large jump in the power signal ofthe remote signal has been detected. Method 700 starts by receiving theincoming receive signal at a repeater which includes the remote signal(step 702). The pilot signal and the receive signal, or blocks of pilotsignal samples and blocks of receive signal samples, are provided to thechannel estimation block for performing channel estimation (step 703).Sub channel estimate for each incoming block of pilot signal samples isthus computed (step 704). Each sub-channel estimate is computed using ablock of N samples of the pilot signal and a block of N samples of thereceive signal.

Then, swings in the power level of the remote signal are detected todetermine if there are large power swings in the remote signal (step705). When no large power swings are detected in the gain control inputsignal (step 706), method 700 returns to step 702 to continue to receivethe incoming receive signal and continue to estimate the feedbackchannel based on the incoming samples of the pilot signal and thereceive signal. However, when a large power swing, such as a power swingof 9 dB or greater, is detected (step 706), one or more sub channelestimates are discarded (step 708). A final channel estimate is computedusing a single undiscarded sub channel estimate or by averaging two ormore undiscarded sub channel estimates (step 710). Method 700 thenreturns to step 702 to continue to receive incoming receive signal.

In one embodiment, both the current and one or more previous sub channelestimates are discarded when a large power swing is detected. When thechannel estimate averaging process is restarted after a large powerswing is detected, the final channel estimate used should be the lastgood final channel estimate, that is, the latest final channel estimatethat does not use the sub channel estimates being discarded. The reasonto discard both the current and the previous channel estimates is thatthere is always some delay between the time a change in remote signalpower occurs, and the time the repeater can detect that change.Therefore, it is possible that a bad sub channel estimate has alreadyoccurred by the time the channel estimation pruning method 700 detectsthe sudden change in receive power. Repeater instability can occur ifonly the current sub channel estimate is discarded. Accordingly, in someembodiments of the channel estimation pruning method 700, both thecurrent and the one or more previous sub channel estimates are discardedwhen a large power swing is detected and the last good final channelestimate is used to continue averaging of the channel estimates. Inother embodiments, method 700 discards all of the sub channel estimatesthat may be corrupted by the power swing so that no corrupted subchannel estimate is retained.

Channel estimation module 260 in repeater 250 may be used to implementthe channel estimation pruning method of the present invention. Channelestimation module 260 may be implemented in software, hardware,firmware, or some combination, and may be configured to perform thefunctions described in the embodiments described herein. In a softwareimplementation, channel estimation module 260 may comprise memory andone or more processors that may be dedicated to the channel estimationmodule 260 or be shared with other features or modules of the repeater.N-sample blocks indicative of the transmit signal may be stored inmemory, as well as N-sample blocks indicative of the receive signal. Theprocessor may access instructions to read the stored data and performchannel estimation on the pilot signal blocks and receive signal blocks.The processor may access instructions to detect changes in the powerlevel of the remote signal. In some embodiments, the processor mayaccess instructions to discard samples or blocks of samples of the pilotand receive signals, while in some embodiments, one or more sub channelestimates may be discarded. Similarly, in a hardware implementation, thevarious operations may be performed using hardware such as a digitalsignal processor (DSP).

The advantage of applying power detection in an interferencecancellation repeater is that stability is maintained regardless of thesignal dynamics of the remote signal. If there were no power detectionto deal with such signal dynamics, the repeater could potentially becomeunstable. The channel estimation pruning method of the present inventionensures that an on-frequency repeater can run robustly in the presenceof large scale signal dynamics of the remote signal. More specifically,the channel estimation pruning method of the present invention ensuresthat channel estimates do not become corrupted due to undesired signaldynamics so that channel estimation and echo cancellation can runrobustly in the presence of changing signal dynamics of the remotesignal.

Those skilled in the art will understand that information and signalsmay be represented using any of a variety of different technologies andtechniques. For example: data, information, signals, bits, symbols,chips, instructions, and commands may be referenced throughout the abovedescription. These may be represented by voltages, currents,electromagnetic waves, magnetic fields or particles, optical fields orparticles, or any combination thereof.

In one or more exemplary embodiments, the functions and processesdescribed may be implemented in hardware, software, firmware, or anycombination thereof. If implemented in software, the functions may bestored on a computer-readable medium. Computer-readable media includescomputer storage media. A storage media may be any available media thatcan be accessed by a computer. By way of example, and not limitation,such computer-readable media can comprise RAM, ROM, EEPROM, CD-ROM orother optical disk storage, magnetic disk storage or other magneticstorage devices, or any other medium that can be used to carry or storedesired program code in the form of instructions or data structures andthat can be accessed by a computer. Disk and disc, as used herein,includes compact disc (CD), laser disc, optical disc, digital versatiledisc (DVD), floppy disk and blu-ray disc where disks usually reproducedata magnetically, while discs reproduce data optically with lasers.Combinations of the above should also be included within the scope ofcomputer-readable media. The term “control logic” used herein applies tosoftware (in which functionality is implemented by instructions storedon a machine-readable medium to be executed using a processor), hardware(in which functionality is implemented using circuitry (such as logicgates), where the circuitry is configured to provide particular outputfor particular input, and firmware (in which functionality isimplemented using re-programmable circuitry), and also applies tocombinations of one or more of software, hardware, and firmware.

For a firmware and/or software implementation, the methodologies may beimplemented with modules (e.g., procedures, functions, and so on) thatperform the functions described herein. Any machine readable mediumtangibly embodying instructions may be used in implementing themethodologies described herein. For example, software codes may bestored in a memory, for example the memory of mobile station or arepeater, and executed by a processor, for example the microprocessor ofmodem. Memory may be implemented within the processor or external to theprocessor. As used herein the term “memory” refers to any type of longterm, short term, volatile, nonvolatile, or other memory and is not tobe limited to any particular type of memory or number of memories, ortype of media upon which memory is stored.

Various modifications to these implementations will be readily apparentto those skilled in the art, and the generic principles defined hereinmay be applied to other implementations without departing from thespirit or scope of the invention. Thus, the present invention is notintended to be limited to the features shown herein but is to beaccorded the widest scope consistent with the principles and novelfeatures disclosed herein.

What is claimed is:
 1. A method for estimating a feedback channel for awireless repeater in a wireless communication system, the wirelessrepeater having a first antenna and a second antenna to receive areceive signal and transmit an amplified signal, the receive signalbeing a sum of a remote signal to be repeated and a feedback signalresulting from the feedback channel between the first and second antennaof the wireless repeater, the method comprising: (a) selecting blocks ofN samples of the amplified signal in a feedback loop of the repeater asa pilot signal; (b) selecting blocks of N samples of the receive signal;(c) estimating an error correction term using a most recent channelestimate; (d) canceling the error correction term from a current blockof the receive signal to generate a corrected block of the receivesignal; and (e) estimating the feedback channel between the firstantenna and the second antenna using frequency domain channel estimationand using a current block of the pilot signal and the corrected block ofthe receive signal, the channel estimate obtained becoming the mostrecent channel estimate.
 2. The method of claim 1, wherein estimating anerror correction term using a most recent channel estimate comprisesestimating the error correction term using the most recent channelestimate and a current block of N samples pilot signal.
 3. The method ofclaim 1, wherein estimating the feedback channel between the firstantenna and the second antenna using frequency domain channel estimationcomprises performing Fast Fourier transform (FFT) operations on thecurrent blocks of the pilot signal and the corrected block of thereceive signal.
 4. The method of claim 1, further comprising: (f)replacing the channel estimate obtained from estimating the feedbackchannel using the corrected block of the receive signal as the mostrecent channel estimate; and (g) repeating the steps of (a) through (f)on the next block of N samples of the pilot signal and the next block ofN samples of the receive signal.
 5. The method of claim 1, furthercomprising: (f) replacing the channel estimate obtained from estimatingthe feedback channel using the corrected block of the receive signal asthe most recent channel estimate; and (g) repeating the steps of (a)through (f) for a fixed number of iterations on the current block of Nsamples of the pilot signal and the current block of N samples of thereceive signal.
 6. The method of claim 5, further comprising: (h)repeating the steps of (a) through (g) on the next block of N samples ofthe pilot signal and the next block of N samples of the receive signal.7. The method of claim 1, wherein canceling the error correction termfrom a current block of the receive signal comprises: canceling theerror correction term from a current block of the receive signal in thetime domain or in the frequency domain.
 8. The method of claim 1,wherein estimating the feedback channel between the first antenna andthe second antenna using frequency domain channel estimation comprises:estimating the feedback channel between the first antenna and the secondantenna using frequency domain channel estimation and using a currentblock of the pilot signal, the corrected block of the receive signal andone or more previously corrected blocks of the receive signal and thecorresponding blocks of the pilot signal, the channel estimate obtainedbecoming the most recent channel estimate.
 9. The method of claim 1,wherein estimating an error correction term using a most recent channelestimate and canceling the error correction term from a current block ofthe receive signal comprise: estimating a channel estimate error termdirectly based on the most recent channel estimate; and canceling thechannel estimate error term from the most recent channel estimate.
 10. Anon-transitory machine-readable medium comprising instructions, which,when executed by a machine, cause the machine to perform operations, theinstructions comprising: (a) selecting blocks of N samples of theamplified signal in a feedback loop of the repeater as a pilot signal;(b) selecting blocks of N samples of the receive signal; (c) estimatingan error correction term using a most recent channel estimate; (d)canceling the error correction term from a current block of the receivesignal to generate a corrected block of the receive signal; and (e)estimating the feedback channel between the first antenna and the secondantenna using frequency domain channel estimation and using a currentblock of the pilot signal and the corrected block of the receive signal,the channel estimate obtained becoming the most recent channel estimate.11. A non-transitory computer-readable medium including program codestored thereon, comprising: program code to select blocks of N samplesof the amplified signal in a feedback loop of the repeater as a pilotsignal; program code to select blocks of N samples of the receivesignal; program code to estimate an error correction term using a mostrecent channel estimate; program code to cancel the error correctionterm from a current block of the receive signal to generate a correctedblock of the receive signal; and program code to estimate the feedbackchannel between the first antenna and the second antenna using frequencydomain channel estimation and using a current block of the pilot signaland the corrected block of the receive signal, the channel estimateobtained becoming the most recent channel estimate.
 12. A wirelessrepeater, comprising: a first antenna and a second antenna to receive areceive signal and transmit an amplified signal, the receive signalbeing a sum of a remote signal to be repeated and a feedback signalresulting from a feedback channel between the first antenna and thesecond antenna; and a channel estimation module configured to estimatean error correction term using a most recent channel estimate and acurrent block of N samples of a signal in a feedback loop of therepeater indicative of the amplified signal as a pilot signal, to cancelthe error correction term from a current block of N samples of thereceive signal to generate a corrected block of the receive signal, andto estimate the feedback channel between the first antenna and thesecond antenna using frequency domain channel estimation and using thecurrent block of the pilot signal and the corrected block of the receivesignal, the channel estimate obtained becoming the most recent channelestimate.
 13. The wireless repeater of claim 12, wherein the channelestimation module is configured to estimate the feedback channel betweenthe first antenna and the second antenna using frequency domain channelestimation by performing Fast Fourier transform (FFT) operations on thecurrent blocks of the pilot signal and the corrected block of thereceive signal.
 14. The wireless repeater of claim 12, wherein thechannel estimation module is further configured to replace the channelestimate obtained from the feedback channel estimation using thecorrected block of the receive signal as the most recent channelestimate, and to repeat estimating the error correction term, cancelingthe error correction term and estimating the feedback channel on thenext block of N samples of the pilot signal and next block of N samplesof the receive signal.
 15. The wireless repeater of claim 12, whereinthe channel estimation module is further configured to replace thechannel estimate obtained from the feedback channel estimation using thecorrected block of the receive signal as the most recent channelestimate, and to repeat for a fixed number of iterations estimating theerror correction term, canceling the error correction term andestimating the feedback channel on the current block of N samples of thepilot signal and the current block of N samples of the receive signal.16. The wireless repeater of claim 15, wherein the channel estimationmodule is further configured to repeat estimating the error correctionterm, canceling the error correction term and estimating the feedbackchannel on the next block of N samples of the pilot signal and nextblock of N samples of the receive signal.
 17. The wireless repeater ofclaim 12, wherein the channel estimation module is configured to cancelthe error correction term from a current block of N samples of thereceive signal in the time domain or in the frequency domain.
 18. Thewireless repeater of claim 12, wherein the channel estimation module isfurther configured to estimate the feedback channel between the firstantenna and the second antenna using frequency domain channel estimationand using a current block of the pilot signal, the corrected block ofthe receive signal and one or more previously corrected blocks of thereceive signal and the corresponding blocks of the pilot signal, thechannel estimate obtained becoming the most recent channel estimate. 19.The wireless repeater of claim 12, wherein the channel estimation moduleis configured to estimate a channel estimate error term directly basedon the most recent channel estimate and is further configured to thechannel estimate error term from the most recent channel estimateinstead of cancelling the error correction term from a current block ofN samples of the receive signal.
 20. A wireless repeater, comprising:first and second means for receiving a receive signal and transmit anamplified signal, the receive signal being a sum of a remote signal tobe repeated and a feedback signal resulting from a feedback channelbetween the first and second means; and third means for estimating anerror correction term using a most recent channel estimate and a currentblock of N samples of a signal in a feedback loop of the repeaterindicative of the amplified signal as a pilot signal, canceling theerror correction term from a current block of N samples of the receivesignal to generate a corrected block of the receive signal, andestimating the feedback channel between the first antenna and the secondantenna using frequency domain channel estimation and using the currentblock of the pilot signal and the corrected block of the receive signal,the channel estimate obtained becoming the most recent channel estimate.21. A method for estimating a feedback channel for a wireless repeaterin a wireless communication system, the wireless repeater having a firstantenna and a second antenna to receive a receive signal and transmit anamplified signal, the receive signal being a sum of a remote signal tobe repeated and a feedback signal resulting from the feedback channelbetween the first and second antenna of the wireless repeater, themethod comprising: (a) selecting blocks of N samples of the amplifiedsignal in a feedback loop of the repeater as a pilot signal; (b)selecting blocks of N samples of the receive signal; (c) estimating achannel estimate error term using a most recent channel estimate; (d)estimating the feedback channel between the first antenna and the secondantenna using frequency domain channel estimation and using a currentblock of the pilot signal and the corresponding block of the receivesignal; (e) processing the channel estimate error term through thefrequency domain channel estimation; and (f) canceling the channelestimate error term from the computed channel estimate, the channelestimate obtained becoming the most recent channel estimate.
 22. Themethod of claim 3, wherein N is a block size of the FFT operations beingperformed on the current blocks of the pilot signal and the correctedblock of the receive signal.
 23. The method of claim 1, wherein usingfrequency domain channel estimation comprises using frequency domainchannel estimation without a cyclic prefix in the amplified signal.